Variable gain amplifier, mixer and quadrature modulator using the same

ABSTRACT

A variable gain amplifier includes a voltage-to-current converter for converting the input voltage to a current, a current amplifier for amplifying the current converted by the voltage-to-current converter, a current-to-voltage converter for converting the current amplified by the current amplifier into a voltage, and a controller for controlling an amplification factor of the current amplifier.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a variable gain amplifier for use in such as a transmission section of, for example, a CDMA wireless terminal, and a mixer and a quadrature modulator that use such a variable gain amplifier.

2. Description of the Related Art

As a code division multiple access (CDMA) scheme has come to be utilized more and more, research and development has been greatly advancing on a radio that accommodates CDMA (hereinafter referred to as “CDMA wireless terminal”). For a transmission section of the CDMA wireless terminal, it is indispensable to conduct transmission power control of 70 dB or more, so that a variable gain amplifier plays a major role in this power control. The variable gain amplifier for use in the transmission section has requirements of (1) low dissipation power, (2) low distortion, and (3) ensuring gain continuity. Specifically, “low dissipation power” means that it is possible to set to low dissipation power when a gain and an output level are low in level, “low distortion” means that a saturation input level remains unchanged even if a gain changes, and “ensuring gain continuity” means that a gain can be changed continuously.

For example, in one of conventional variable gain amplifiers, a gain is controlled by a gain control circuit having a plurality of differential amplifiers (see Japanese Patent Application Laid Open No. 2000-232328, for example). These differential amplifiers are each made of two transistors in such a configuration that a base of one of the two transistors and that of the counterpart transistor are commonly connected and bases of the other transistors are connected to different reference voltage terminals. Further, collectors of the two transistors of each of the differential amplifiers are commonly connected and a load resistor of each of the transistors is constituted of a resistor element. The commonly-connected base of one of the transistors of each of the differential amplifiers is supplied with a voltage proportional to a thermal voltage which is obtained by converting an externally applied first gain-control voltage. An output voltage of each of the differential amplifiers is supplied via a buffer to the variable gain amplifier as a second gain-control voltage. The variable gain amplifier in which a gain is controlled by such a gain control circuit has a feature of ensuring continuity of a gain with respect to an externally applied gain control voltage with its relative simple configuration and improving linearity of that gain. Hereinafter, the term “gain control voltage” means a gain control signal in the description.

Another conventional variable gain amplifier has a first differential amplification circuit constituted of a first couple of transistors and a second differential amplification circuit constituted of a second couple of transistors (see Japanese Patent Application Laid Open No. 2003-023331, for example). In this variable gain amplifier, when an externally applied gain-control voltage increases, a bias current of the first differential amplifier having a larger gain increases, so that the gain of the first differential amplifier is increased, thereby increasing an overall gain. If the externally applied gain control voltage decreases, on the other hand, a bias current of the second differential amplifier having a smaller gain increases, so that the gain of the second differential amplifier is increased and hence its influence is increased, thus resulting in an amplification circuit having a higher saturation input level and a lower gain. In this configuration, as a gain increases, a bias current of the second differential amplifier decreases, so that dissipation power can be reduced. That is, this variable gain amplifier features that dissipation power can be reduced.

In a further conventional variable gain amplifier, when converting an input voltage into a current and amplifying this converted current to output the resulting current, a quantity of the output current is controlled by changeover of switches in accordance with an externally applied gain control voltage (see Japanese Patent Application Laid Open No. H11-340760, for example). This variable gain amplifier has a feature that a dissipation current can be reduced if a gain is low.

Furthermore, in some other conventional variable gain amplifiers, a bias current generation circuit that supplies a bias current to a linear amplifier constituting a communication high-frequency power amplification circuit is comprised of a plurality of variable-current sources having different current values at starting levels, and these variable current sources are controlled by using an input control voltage. Then, the resultant controlled currents may be combined into a bias current and this combined current may change exponentially with respect to the input control voltage (see Japanese Patent Application Laid Open No. 2003-218649, for example).

A still further conventional variable gain amplifier is comprised of a mutual conductance amplifier having a degenerate impedance element and a variable-current source that is arranged to supply a current to this mutual conductance amplifier and that is configured so that the current may change simultaneously with an effective degenerate impedance of the mutual conductance amplifier, in accordance with a gain control voltage (V_(GC)) (see Japanese Patent Application Laid Open No. 2003-229735, for example).

However, a variable gain amplifier in which a gain is controlled by a gain control circuit having a plurality of differential amplifiers has a disadvantage that a dissipation current cannot be changed for each of the gains. Another variable gain amplifier having two differential amplifier has a disadvantage that, although a saturation input level is low at most of intermediate level gains, the saturation input level increases if an influence of the first differential amplification circuit is reduced due to an increase in bias current of the second differential amplification circuit having a smaller gain, that is, the saturation input level fluctuates. A further variable gain amplifier for controlling a quantity of a current by changeover of switches has a disadvantage that it can conduct only discrete gain control on an externally applied gain control voltage but cannot conduct continuous control thereon.

That is, the conventional variable gain amplifiers have a problem that they cannot satisfy the above-described three features of: (1) low dissipation power that low dissipation power can be set if a gain is low and hence an output level is low, (2) low distortion that a saturation input level remains unchanged even if a gain changes, and (3) gain continuity that a continuous gain can be ensured.

SUMMARY OF THE INVENTION

The present invention has been made to solve the above-described problems, and aims to provide a variable gain amplifier that satisfies the three features of lower power dissipation, low distortion, and gain continuity simultaneously, and to provide a mixer and a quadrature modulator that utilize such a variable gain amplifier.

A variable gain amplifier according to the present invention amplifies an input voltage at a variable gain and outputs the amplified voltage. The variable gain amplifier includes a voltage-to-current converter for converting the input voltage into a current; a current amplifier for amplifying the current converted by the voltage-to-current converter; a current-to-voltage converter for converting the current amplified by the current amplifier into a voltage and outputting the resulting voltage; and a controller for controlling the gain of the variable gain amplifier by varying an amplification factor of the current amplifier in accordance with a gain control signal. Hereinafter, this variable gain amplifier is referred to as a “first variable gain amplifier”.

Preferably, in the first variable gain amplifier, the current amplifier has a current mirror circuit to which a current converted by the voltage-to-current converter is inputted, the controller varies a ratio of an output current to an input current of the current mirror circuit in accordance with the gain control signal, and the current-to-voltage converter converts an output current of the current mirror circuit into a voltage and outputs the resulting voltage. Hereinafter, this variable gain amplifier is referred to as a “second variable gain amplifier”.

Preferably, in the second variable gain amplifier, the current mirror circuit includes an input-side transistor through which the current converted by the voltage-to-current converter flows; an output-side transistor through which a current proportional to the current flowing through the input-side transistor flows; and a variable resistor connected in series to the output-side transistor. The current-to-voltage converter converts the current flowing through the output-side transistor into a voltage and outputs the resulting voltage, and the controller varies a resistance of the variable resistor in accordance with the gain control signal. Hereinafter, this variable gain amplifier is referred to as a “third variable gain amplifier”.

Preferably, in the third variable gain amplifier, the variable resistor is constituted of an MOS transistor, and the controller varies a control voltage of the MOS transistor in accordance with the gain control signal. Hereinafter, this variable gain amplifier is referred to as a “fourth variable gain amplifier”.

Preferably, in the first variable gain amplifier, the current amplifier has a plurality of current amplification circuits each amplifying the current converted by the voltage-to-current converter and outputting the amplified current, the controller varies an amplification factor of each of the current amplification circuits in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents outputted from the current amplification circuits into a voltage. Hereinafter, this variable gain amplifier is referred to as a “fifth variable gain amplifier”.

Preferably, in the fifth variable gain amplifier, when N and n are integers equal to or larger than 2, α₀and A are arbitrary constants, and a relation of 1<n<N+1 is established, given that the number of the current amplification circuits is N, a maximum current amplification factor N_(n) of the n-th current amplification circuit is α₀×(A−1)×A^((n−2)). Hereinafter, this variable gain amplifier is referred to as a “sixth variable gain amplifier”.

Preferably, in the fifth or sixth variable gain amplifier, the voltage-to-current converter has a pair of first and second transistors through each of which a current converted from the input voltage flows, each of the current amplification circuits has a pair of third and fourth transistors to constitute a group of third transistors and a group of fourth transistors in the current amplifier. Base electrodes of the group of the third transistors are commonly connected to the base of the first transistor and collector electrodes of the group of the third transistors are commonly connected so that a current proportional to a current flowing through the first transistor flows through each of the group of the third transistors. Base electrodes of the group of the fourth transistors are commonly connected to the base of the second transistor and collector electrodes of the group of the fourth transistors are commonly connected so that a current proportional to a current flowing through the second transistor flows through each of the group of the fourth transistors. The controller varies a quantity of a current flowing through each of the third and fourth transistors in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents flowing through the group of the third transistors and the group of the fourth transistors into a voltage. Hereinafter, this variable gain amplifier is referred to as a “seventh variable gain amplifier”.

Preferably, in the seventh variable gain amplifier, each of the current amplification circuits has a variable resistor connected in series to each of the third transistors and a variable resistor connected in series to each of the fourth transistors, and the controller varies a resistance of each of the variable resistors in accordance with the gain control signal. Hereinafter, this variable gain amplifier is referred to as an “eighth variable gain amplifier”.

Preferably, in the seventh variable gain amplifier, each of the current amplification circuits has a MOS transistor connected in series to each of the third transistors and a MOS transistor connected in series to each of the fourth transistors, and the controller varies a control voltage of each of the MOS transistors in accordance with the gain control signal, thereby varying a quantity of a current flowing through each of the third and fourth transistors. Hereinafter, this variable gain amplifier is referred to as a “ninth variable gain amplifier”.

Preferably, in the ninth variable gain amplifier, the controller has a constant voltage generator that generates a constant voltage to be applied to each of the current amplification circuits and a plurality of control voltage generators each of which generates and outputs a control voltage to be applied to each pair of the MOS transistors in each of the current amplification circuits. Each of the control voltage generators has a differential amplification circuit to which a constant voltage generated by the constant voltage generator and a voltage of the gain control signal are inputted to the input terminals thereof, respectively, a current mirror circuit whose input-side transistor constitutes a load of one of a couple of differential transistors of the differential amplification circuit, and a voltage output section for converting an output current of the current mirror circuit into a voltage and outputting the resultant converted voltage. Hereinafter, this variable gain amplifier is referred to as a “tenth variable gain amplifier”.

Preferably, in any one of the first through tenth variable gain amplifiers, the current-to-voltage converter may be provided with a resistor.

Preferably, in any one of the first through tenth variable gain amplifiers, the current-to-voltage converter may be provided with an inductor.

Preferably, in any one of the first through tenth variable gain amplifiers, the current-to-voltage converter may be provided with an inductor and a capacitor connected in parallel to each other.

A mixer according to the present invention converts a frequency of an input voltage in accordance with a conversion gain control signal supplied from an outside and outputs the resulting voltage having a frequency converted. The mixer includes a voltage-to-current converter for converting the input voltage into a current; a current amplifier for amplifying the current converted by the voltage-to-current converter; a controller for controlling an amplification factor of the current amplifier in accordance with a gain control signal; a mixing unit for converting the frequency of the current amplified by the current amplifier, in accordance with the conversion gain control signal; and a current-to-voltage converter for converting the resulting frequency-converted current into a voltage. Hereinafter, this mixer is referred to as a “first mixer”.

Preferably, in the first mixer, the current amplifier has a current mirror circuit to which a current converted by the voltage-to-current converter is inputted, the controller varies a ratio of an output current to an input current of the current mirror circuit in accordance with the gain control signal, and the current-to-voltage converter converts an output current of the current mirror circuit into a voltage and outputs the resulting voltage. Hereinafter, this mixer is referred to as a “second mixer”.

Preferably, in the second mixer, the current mirror circuit includes an input-side transistor through which the current converted by the voltage-to-current converter flows; an output-side transistor through which a current proportional to the current flowing through the input-side transistor flows; and a variable resistor connected in series to the output-side transistor. The current-to-voltage converter converts the current flowing through the output-side transistor into a voltage and outputs the resulting voltage, and the controller varies a resistance of the variable resistor in accordance with the gain control signal. Hereinafter, this mixer is referred to as a “third mixer”.

Preferably, in the third mixer, the variable resistor is constituted of a MOS transistor and the controller varies a control voltage of the MOS transistor in accordance with the gain control signal. Hereinafter, this mixer is referred to as a “fourth mixer”.

Preferably, in the first mixer, the current amplifier has a plurality of current amplification circuits each amplifying the current converted by the voltage-to-current converter and outputting the amplified current, the controller varies an amplification factor of each of the current amplification circuits in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents outputted from the current amplification circuits into a voltage. Hereinafter, this mixer is referred to as a “fifth mixer”.

Preferably, in the fifth mixer, when N and n are integers equal to or larger than 2, α₀ and A are arbitrary constants, and a relation of 1<n<N+1 is established, given that the number of the current amplification circuits is N, a maximum current amplification factor N_(n) of the n-th current amplification circuit is α₀×(A−1)×A^((n−2)). Hereinafter, this mixer is referred to as a “sixth mixer”.

Preferably, in the fifth or sixth mixer, the voltage-to-current converter has a pair of first and second transistors through each of which a current converted from the input voltage flows, each of the current amplification circuits has a pair of third and fourth transistors to constitute a group of third transistors and a group of fourth transistors in the current amplifier. Base electrodes of the group of the third transistors are commonly connected to the base of the first transistor and collector electrodes of the group of the third transistors are commonly connected so that a current proportional to a current flowing through the first transistor flows through each of the group of the third transistors. Base electrodes of the group of the fourth transistors are commonly connected to the base of the second transistor and collector electrodes of the group of the fourth transistors are commonly connected so that a current proportional to a current flowing through the second transistor flows through each of the group of the fourth transistors. The controller varies a quantity of a current flowing through each of the third and fourth transistors in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents flowing through the group of the third transistors and the group of the fourth transistors into a voltage. Hereinafter, this mixer is referred to as a “seventh mixer”.

Preferably, in the seventh mixer, each of the current amplification circuits has a variable resistor connected in series to each of the third transistors and a variable resistor connected in series to each of the fourth transistors, and the controller varies a resistance of each of the variable resistors in accordance with the gain control signal. Hereinafter, this mixer is referred to as an “eighth mixer”.

Preferably, in the seventh mixer, each of the current amplification circuits has a MOS transistor connected in series to each of the third transistors and a MOS transistor connected in series to each of the fourth transistors, and the controller varies a control voltage of each of the MOS transistors in accordance with the gain control signal, thereby varying a quantity of a current flowing through each of the third and fourth transistors. Hereinafter, this mixer is referred to as a “ninth mixer”.

Preferably, in the ninth mixer, the controller has a constant voltage generator that generates a constant voltage to be applied to each of the current amplification circuits and a plurality of control voltage generators each of which generates and outputs a control voltage to be applied to each pair of the MOS transistors in each of the current amplification circuits. Each of the control voltage generators has a differential amplification circuit to which a constant voltage generated by the constant voltage generator and a voltage of the gain control signal are inputted to the input terminals thereof, respectively, a current mirror circuit whose input-side transistor constitutes a load of one of a couple of differential transistors of the differential amplification circuit, and a voltage output section for converting an output current of the current mirror circuit into a voltage and outputting the resultant converted voltage. Hereinafter, this mixer is referred to as a “tenth mixer”.

Preferably, in any one of the first through tenth mixers, the current-to-voltage converter may be provided with a resistor.

Preferably, in any one of the first through tenth mixers, the current-to-voltage converter may be provided with an inductor.

Preferably, in any one of the first through tenth mixers, the current-to-voltage converter may be provided with an inductor and a capacitor connected in parallel to each other.

A quadrature modulator according to the present invention includes a pair of mixers, an adder for adding up output currents of the mixers, and a current-to-voltage converter for converting an output current of the adder into a voltage. Each of the mixers has a voltage-to-current converter for converting an input voltage into a current, a current amplifier for amplifying the current converted by the voltage-to-current converter, a controller for controlling an amplification factor of the current amplifier in accordance with a gain control signal, and a mixing unit for converting a frequency of the current amplified by the current amplifier, in accordance with a conversion gain control signal.

A variable gain amplifier according to the present invention includes a voltage-to-current converter for converting an input voltage into a current, a current amplifier for amplifying the current, a current-to-voltage converter for converting the current amplified by the current amplifier into a voltage, and a controller for controlling an amplification factor of the current amplifier in accordance with a gain control signal. Therefore, the variable gain amplifier can satisfy the three features of low dissipation power, low distortion, and gain continuity simultaneously.

A mixer according to the present invention includes a voltage-to-current converter for converting an input voltage into a current, a current amplifier for amplifying the current converted by the voltage-to-current converter, a controller for controlling an amplification factor of the current amplifier in accordance with a gain control signal externally inputted, a mixing unit for converting a frequency of the current amplified by the current amplifier, and a current-to-voltage converter for converting the current with its frequency converted by the mixing unit into a voltage. Therefore, the mixer can satisfy the three features of low dissipation power, low distortion, and gain continuity simultaneously.

A quadrature modulator according to the present invention includes a pair of mixers, an addition section for adding up output currents of the mixers, and a current-to-voltage converter for converting an output current of the addition section into a voltage, wherein each mixer has a voltage-to-current converter for converting an input voltage into a current, a current amplifier for amplifying the current converted by the voltage-to-current converter, a controller for controlling an amplification factor of the current amplifier in accordance with a gain control signal, and a mixing unit for converting a frequency of the current amplified by the current amplifier, in accordance with a gain control signal. Therefore, the quadrature modulator can satisfy the three features of low dissipation power, low distortion, and gain continuity simultaneously.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram showing a configuration example of a variable gain amplifier according to a first embodiment of the present invention;

FIG. 2 is a circuit diagram showing a configuration example of a variable gain amplifier according to a second embodiment of the present invention;

FIG. 3 is a graph showing a relationship between an input gain control voltage V_(GC) and voltages generated at output terminals S1-S4;

FIG. 4 is a circuit diagram showing a configuration example of a control voltage generator 15;

FIG. 5 is a circuit diagram showing a configuration example of a voltage comparison circuit 42;

FIG. 6A is a graph showing a relationship between the gain control voltage V_(GC) and voltages V₅₁ and V₅₂ to the corresponding input terminals 51 and 52, respectively;

FIG. 6B is a graph showing a relationship between the gain control voltage G_(GC) and a voltage output to an output terminal 59;

FIG. 7 is a graph showing a relationship between the gain control voltage G_(GC) and a gain G_(v) for the variable gain amplifier according to the first embodiment;

FIG. 8 is a graph showing a relationship between the gain control voltage G_(GC) and the gain G_(v) in a case where the number of a current amplification circuits is set to 10;

FIG. 9 is a graph showing a relationship between the gain control voltage V_(GC) and a current i₁₁;

FIG. 10 is a circuit diagram showing a configuration example in the case where a current-to-voltage converter 14 is constituted of an inductor;

FIG. 11 is a graph showing a relationship between a voltage V_(o1) at an output terminal 19 and a current i₁₁ flowing through a resistor R31 in the case where the current-to-voltage converter 14 is constituted of the resistors R31 and R32;

FIG. 12 is a graph showing a relationship between the voltage V_(o1) at the output terminal 19 and the current i₁₁ flowing through an inductor L31 in the case where the current-to-voltage converter 14 is constituted of the inductors L31 and L32;

FIG. 13 is a circuit diagram showing a configuration example in the case where the current-to-voltage converter 14 is constituted of an inductor and a capacitor connected in parallel to each other;

FIG. 14 is a circuit diagram showing another configuration example of a voltage-to-current converter 12;

FIG. 15 is a circuit diagram showing a mixer that uses the variable gain amplifier 11 according to the second embodiment of the present invention; and

FIG. 16 shows a circuit diagram of a quadrature modulator that uses the variable gain amplifier 11 according to the second embodiment of the present invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The following will describe embodiments of the present invention with reference to the drawings.

First Embodiment

FIG. 1 is a circuit diagram showing a configuration example of a variable gain amplifier according to the first embodiment of the present invention. This variable gain amplifier amplifies an input voltage at a variable gain and outputs the resultant voltage. As shown in FIG. 1, the variable gain amplifier 1 includes a voltage-to-current converter 2, a current amplifier 3, a current-to-voltage converter 4, and a controller 5. The voltage-to-current converter 2 converts an input voltage into a current. The current amplifier 3 amplifies the current converted by the voltage-to-current converter 2. The current-to-voltage converter 4 converts the current amplified by the current amplifier 3 into a voltage. The controller 5 changes an amplification factor of the current amplifier 3 in accordance with a gain control signal inputted from an outside.

The following describes in more detail the configuration of the variable gain amplifier 1. The voltage-to-current converter 2 includes a resistor R1 connected to a power source voltage V_(cc), and a PNP transistor Q1 connected in series to the resistor R1. The current amplifier 3 includes a pair of NPN transistors Q2 and Q3 that constitute a current mirror circuit and resistors R2 and R3 connected in series to the NPN transistors Q2 and Q3, respectively. In this arrangement, the NPN transistor Q2 is connected in series to the PNP transistor Q1, and collectors of the PNP transistor Q1 and the NPN transistor Q2 are connected to each other, and the resistor R3 is a variable resistor. The current-to-voltage converter 4 includes a resistor R4 whose one terminal is connected to the power source voltage V_(cc). The other terminal of the resistor R4 is connected to a collector of the NPN transistor Q3. The controller 5 changes a resistance of the variable resistor R3 in accordance with a gain control signal inputted from an outside.

The following describes an operation of the variable gain amplifier 1. An input terminal T1 is connected to a gate of the NPN transistor Q1 in the voltage-to-current converter 2. When a voltage V_(i) is applied to the input terminal T1, a current i_(n) flows through the PNP transistor Q1. In this arrangement, a conversion factor gm of the voltage-to-current converter 2 is given by a following Equation (1) using a resistance r_(i) of the resistor R1: gm=Δi _(a) /ΔV _(i)=1/r ₁  (1)

Since the PNP transistor Q1 and the NPN transistor Q2 are connected in series to each other, the current ia flows also through the NPN transistor Q2. When the current i_(a) flows through the NPN transistor Q2, a current i_(b) flows also through the NPN transistor Q3 according to characteristics of the current mirror circuit. In this case, supposing that base-emitter voltages of the NPN transistors Q2 and Q3 are V_(BE2) and V_(BE3) respectively and resistances of the resistors R2 and R3 are r₂ and r₃ respectively, the following Equation (2) is established: r ₂ ×i _(a) +V _(BE2) =r ₃ ×i _(b) +V _(BE3)  (2)

By differentiating both sides of Equation (2) with i_(a), the following Equation (3) is established: r ₂ +ΔV _(BE2) /Δi _(a) =r ₃ ×Δi _(b) /Δi _(a) +ΔV _(BE3) /Δi _(a)  (3)

In this case, if a current amplification factor of the current amplifier 3 is supposed to be G1=Δi_(b)/Δi_(a), the following Equation (4) is established: G1={r ₂ +ΔV _(BE2) /Δi _(a) −ΔV _(BE3) /Δi _(a)}/r₃  (4)

Since r₂>>ΔV_(BE2)/Δi_(a)−ΔV_(BE3)/Δi_(a), the following Equation (5) is established: G1=r ₂ /r ₃  (5)

Further, the current i_(b) flows also through the resistor R4 in the current-to-voltage converter 4. An output terminal T2 is connected to the resistor R4, and a voltage V_(o) proportional to the current i_(b) is outputted from the terminal T2. If a value of the power source voltage is supposed to be V_(cc) and a resistance of the resistor R4 is supposed to be R_(4o), the voltage V_(o) is given as V_(o)=V_(cc)−R_(4o)×i_(b), so that a gain of the variable gain amplifier G_(v)=ΔV_(o)/ΔV_(i) is given by the following Equation (6): $\begin{matrix} \begin{matrix} {G_{v} = {{{\Delta\left( {V_{cc} - {R_{40} \times G\quad 1 \times i_{a}}} \right)}/\Delta}\quad V_{i}}} \\ {= {{- R_{40}} \times G\quad 1 \times \left( {\Delta\quad{i_{a}/\Delta}\quad V_{i}} \right)}} \\ {= {{- R_{40}} \times G\quad 1 \times {gm}}} \\ {= {{- \left( {R_{40}/r_{1}} \right)} \times \left( {r_{2}/r_{3}} \right)}} \end{matrix} & (6) \end{matrix}$

It is noted here that only the resistance r₃ is variable and the resistances r₁, r₂, and R_(4o) are constant. Therefore, it can be known that by changing the resistance r₃ continuously, it is possible to continuously control the gain G_(v) of the variable gain amplifier 1. This can be realized in a manner such that the controller 5 continuously changes the resistance r₃ of the resistor R3 in accordance with the continuously changing gain control signal inputted to a terminal T3.

Further, it can be known that in the variable gain amplifier 1 shown in FIG. 1, if the gain G_(v) is high, the resistance r₃ is small and so the current i _(b) is large, and if the gain G_(v) is low, the resistance r₃ is large and so the current i_(b) is small. That is, according to the variable gain amplifier 1 shown in FIG. 1, if the gain is low and an output level is low, a feature of low dissipation power can be satisfied. Further, the variable gain amplifier 1 can satisfy a feature of low distortion because its saturation input level does not change even if the gain G_(v) changes.

In the variable gain amplifier 1 shown in FIG. 1, although the voltage-to-current converter 2 is constituted of the resistor R1 and the PNP transistor Q1, the present invention is not limited to this. However, preferably the voltage-to-current converter 2 may be configured so as to have a constant conversion rate at which a voltage is converted into a current.

Second Embodiment

FIG. 2 is a circuit diagram showing a configuration example of a variable gain amplifier according to the second embodiment of the present invention. As shown in FIG. 2, a variable gain amplifier 11 according to the second embodiment includes a voltage-to-current converter 12, a current amplifier 13, a current-to-voltage converter 14, and a control voltage generator 15. The voltage-to-current converter 12 includes two input terminals 16 and 17 to which input signals are inputted respectively from an outside, and the control voltage generator 15 includes one input terminal 18 to which a gain control signal is inputted from an outside. The current-to-voltage converter 14 converts currents outputted from the current amplifier 13 into voltages and generates the resultant voltages through two output terminals 19 and 20 thereof, respectively. In this arrangement, the control voltage generator 15 constitutes a controller. This variable gain amplifier 11 amplifies the voltages inputted to the input terminals 16 and 17, at the respective variable gains, and outputs them from the corresponding output terminals 19 and 20. That is, the variable gain amplifier 11 shown in FIG. 2 is configured to combine two variable gain amplifiers 1 shown in FIG. 1.

The voltage-to-current converter 12 includes operational amplifiers 21 and 22, constant current sources 23 and 24, NPN transistors Q110 and Q120, resistors R10, R110, and R120, and N type metal-oxide-semiconductor transistors (hereinafter referred to as “N type MOS transistors”) M110 and M120. The current amplifier 13 includes four current amplification circuits 31, 32, 33, and 34. The current amplification circuit 31 includes NPN transistors Q111 and Q121, resistors R111 and R121, and N type MOS transistors M111 and M121, and the current amplification circuit 32 includes NPN transistors Q112 and Q122, resistors R112 and R122, and N type MOS transistors M112 and M122. Further, the current amplification circuit 33 includes NPN transistors Q113 and Q123, resistors R113 and R123, and N type MOS transistors M113 and M123, and the current amplification circuit 34 includes NPN transistors Q114 and Q124, resistors R114 and R124, and N type MOS transistors M114 and M124. The current-to-voltage converter 14 includes a pair of resistors R31 and R32 connected to a power source voltage V_(cc).

To simplify description, in the following, it is supposed that the NPN transistors Q110 to Q114 and Q120 to Q124 have the same size and the same properties and the resistors R110 to R114 and R120 to R124 have the same resistance. Further, it is supposed that the N type MOS transistors M110 to M114 and M120 to M124 have the same size and properties and the resistors R31 and R32 have the same resistance.

The control voltage generator 15 includes five output terminals S0-S4. The output terminal S0 is connected to each of gates of the N type MOS transistors M110 and M120 in the voltage-to-current converter 12, and the output terminal S1 is connected to each of gates of the N type MOS transistors M111 and M121 in the current amplification circuit 31. Further, the output terminal S2 is connected to each of gates of the N type MOS transistors M112 and M122 in the current amplification circuit 32, the output terminal S3 is connected to each of gates of the N type MOS transistors M113 and M123 in the current amplification circuit 33, and the output terminal S4 is connected to each of gates of the N type MOS transistors M114 and M124 in the current amplification circuit 34. When a gain control voltage VGC serving as is a gain control signal is inputted to the input terminal 18 from the outside, the control voltage generator 15 generates voltages in accordance with the gain control voltage V_(GC) at the output terminals S0-S4, respectively.

FIG. 3 is a graph showing a relationship between the input gain control voltage V_(GC) and the voltages generated at the output terminals S1-S4. It is supposed here that voltages at which the N type MOS transistors M110-M114 and M120-M124 are turned on and off are, respectively, V_(on) and V_(off). As shown in FIG. 3, if the gain control voltage V_(GC) exceeds a predetermined voltage V_(b), a voltage V_(S1) that appears at the output terminal S1 rapidly increases to exceed the voltage V_(on). Similarly, if the gain control voltage V_(GC) exceeds predetermined voltages V_(c), V_(d), and V_(e), corresponding voltages V_(S2) to V_(S4) that appear at the respective output terminals S2 to S4 also increase rapidly to exceed the voltage V_(on). It is to be noted that the values V_(b), V_(c), V_(d), and V_(e) of the voltage V_(GC) at which the voltages appearing at the respective output terminals S1-S4 change rapidly are set to be different for each of the output terminals S1, S2, S3, and S4. Further, a voltage V_(S0) that appears at the output terminal S0 is set to be always larger than the voltage V_(on) irrespective of a value of the gain control voltage V_(GC).

FIG. 4 is a circuit diagram showing a configuration example of the control voltage generator 15. As shown in FIG. 4, the control voltage generator 15 includes a reference voltage generation circuit 41 and four voltage comparison circuits 42-45. The reference voltage generation circuit 41 has a configuration in which a current source 46 and four resistors R47 to R50 are connected in series between a power source voltage V_(cc) and a ground voltage. The voltage comparison circuit 42 compares a voltage of a node between the resistor R47 and the current source 46 and the gain control voltage V_(GC) inputted through the input terminal 18 and outputs the voltage V_(S1) based on a result of the comparison. The voltage comparison circuit 43 compares a voltage of a node between the resistors R47 and R48 and the gain control voltage V_(GC) and outputs the voltage V_(S2) based on a result of the comparison. The voltage comparison circuit 44 compares a voltage of a node between the resistors R48 and R49 and the gain control voltage V_(GC) and outputs the voltage V_(S3) based on a result of the comparison. The voltage comparison circuit 45 compares a voltage of a node between the resistors R49 and R50 and the gain control voltage V_(GC) and outputs the voltage V_(S4) based on a result of the comparison. In this configuration, the reference voltage generation circuit 41 constitutes a constant voltage generator and each of the voltage comparison circuits 42-45 constitutes a control voltage generation section. For simplicity, it is assumed in the following description that the resistors R47-R50 have the same resistance and the voltage comparison circuits 42-45 have the same configuration.

FIG. 5 is a circuit diagram showing a configuration example of the voltage comparison circuit 42. As shown in FIG. 5, the voltage comparison circuit 42 includes two input terminals 51 and 52, one output terminal 59, a pair of NPN transistors Q141 and Q142, a pair of PNP transistors Q143 and Q144, resistors R141-R145 and a capacitor C140. In this configuration, the NPN transistors Q141 and Q142 and the resistors R141 and 142 constitute a differential amplification circuit and the PNP transistors Q143 and Q144 and the resistors R143 and R144 constitute a current mirror circuit. In this arrangement, the PNP transistor Q143 and the resistor R143 in the current mirror circuit constitute a load of the NPN transistor Q142 which is one of the differential pair of the differential amplification circuit. The resistor R145 constitutes a voltage output section, which converts an output current of the current mirror circuit into a voltage to be outputted. A voltage that appears at the output terminal 59 can be adjusted by adjusting a current from a current source 150, a resistance of the resistor R145, and a rate of an output current with respect to an input current for the current mirror circuit. The capacitor C140 connected to the output terminal 59 is provided to attenuate a noise that appears at the output terminal 59.

The following describes an operation of the voltage comparison circuit 42. FIG. 6A is a graph showing a relationship between the gain control voltage V_(GC) and voltages V₅₁ and V₅₂ inputted to the corresponding input terminals 51 and 52, respectively. FIG. 6B is a graph showing a relationship between the gain control voltage G_(GC) and a voltage outputted to the output terminal 59. As shown in FIG. 6A, the voltage comparison circuit 42 is supplied with a constant voltage at its input terminal 52 from the reference voltage generation circuit 41, and is supplied with the gain control voltage V_(GC) at its input terminal 51. When such voltages V₅₁ and V₅₂ as shown in FIG. 6A are inputted to the corresponding input terminals 51 and 52, a voltage such as shown in FIG. 6 b appears at the output terminal 59. In the voltage comparison circuit 42 shown in FIG. 5, if the two input voltages V₅₁ and V₅₂ are equal to each other, the same current flows through the NPN transistors Q141 and Q142. A current flowing through the PNP transistor Q143 is equal to that flowing through NPN transistor Q142. Further, according to properties of the current mirror circuit, a current flowing through the PNP transistor Q144 is proportional to that flowing through the PNP transistor Q143. A current flowing through the PNP transistor Q144 flows through the resistor R145. In this case, preferably a voltage that appears at the output terminal 59 is adjusted to a threshold value at which N type MOS transistors M111 and M121 are turned on. As shown in FIG. 6B, supposing this threshold voltage to be V_(on), if the input voltage V₅₁ is larger than the input voltage V₅₂, the voltage at the output terminal 59 is larger than V_(on), and if the input voltage V₅₂ is larger than the input voltage V₅₁, the voltage at the output terminal 59 is smaller than V_(on). This holds true also with the voltage comparison circuits 43 to 45. In such a manner, the voltage comparison circuits 42 to 45 can be configured relatively easily.

The following describes an operation of the variable gain amplifier 11 shown in FIG. 2. To simplify the following description, it is supposed that, as for values V_(b), V_(c), V_(d), V_(e) (see FIG. 3) of the gain control voltage V_(GC), voltage differences V_(b)−V_(c), V_(c)−V_(d), and V_(d)−V_(e) are equal to each other. First, it is supposed that the gain control voltage V_(GC) is sufficiently large and the voltages that appear at the output terminals S1, S2, S3, and S4 of the control voltage generation circuit 15 respectively are larger than V_(on). As described above, the gates of the N type MOS transistors M110-M114 and M120-M124 are connected to the control terminals S0-S4 respectively and the voltages larger than V_(on) are generated at the control terminals S0-S4, so that the N type MOS transistors M110-M114 and M120-M124 are all in the on-state. It is to be noted that the N type MOS transistors M110 and M120 are inserted in consideration of fluctuations in turn-on resistances of the N type MOS transistors M111-M114 and M121-M124.

In a case where the N type MOS transistors M110-M114 and M120-M124 are in the on-state as described above, if voltages V_(i1) and V_(i2) are inputted to the input terminals 16 and 17 of the voltage-to-current converter 12 respectively, the voltage V_(i1) is inputted to a non-inverting input terminal (+) of the operational amplifier 21, and an output voltage of the operational amplifier 21 is inputted to a base of the NPN transistor Q110. A collector of the NPN transistor Q110 is connected in common to the constant current source 23 and the resistor R10. Further, a collector voltage of the NPN transistor Q110 is fed back to an inverting input terminal (−) of the operational amplifier 21. An emitter of the NPN transistor Q110 is grounded via the resistor R110 and the N type MOS transistor M110, thereby constituting a grounded-emitter amplifier. The collector voltage of the NPN transistor Q110, which is an output voltage of this grounded-emitter amplifier, is exposed to feedback control by the operational amplifier 21 so as to be equal to the voltage V_(i1) inputted to the input terminal 16.

On the other hand, the voltage V_(i2) is inputted to the non-inverting input terminal (+) of the operational amplifier 22, and an output voltage of the operational amplifier 22 is inputted to a base of the NPN transistor Q120. A collector of the NPN transistor Qi20 is connected in common to the constant current source 24 and the resistor R10. Further, a collector voltage of the NPN transistor Q120 is fed back to an inverting input terminal (−) of the operational amplifier 22. An emitter of the NPN transistor Q120 is grounded via the resistor R120 and the N type MOS transistor M120, thereby constituting a grounded-emitter amplifier. The collector voltage of the NPN transistor Q120, which is an output voltage of this grounded-emitter amplifier, is exposed to feedback control by the operational amplifier 22 so as to be equal to the voltage V_(i2) inputted to the input terminal 17.

If it is supposed that ΔV_(i)=V_(i1)−V_(i2) and that a resistance of the resistor R10 is r₁₀, a current Δi flowing through the resistor R10 is given by the following Equation (7): Δi=(V _(i1) −V _(i2))/r ₁₀ =ΔV _(i) /r ₁₀  (7)

Further, if output currents from the constant current sources 23 and 24 are both supposed to be i₀, a current i₁₁₀ flowing through the NPN transistor Q110 and a current i₁₂₀ flowing through the NPN transistor Q120 are given by the following Equations (8) and (9) respectively: i ₁₁₀ =i ₀ −Δi=i ₀ −ΔV _(i) /r ₁₀  (8) i ₁₂₀ =i ₀ +Δi=i ₀ +ΔV _(i) /r ₁₀  (9)

In this case, relationships of i₁₁₀>0 and i₁₂₀>0 are established, Δi changes in a range of −i₀ through +i₀ in accordance with ΔV_(i). A voltage of ΔV_(i) required for changing Δi in the range of −i₀ through +i₀ is r₁₀×i₀. Conversely, even if a larger voltage of ΔV_(i) is inputted, Δi cannot follow a changing value of ΔV_(i). Therefore, the voltage of r₁₀×i₀ provides a saturation input level of the variable gain amplifier 11.

Next, referring to the NPN transistor Q111, the resistor R111, and the MOS transistor M111 in the current amplification circuit 31 of the current amplifier 13, in a similar manner to the NPN transistor Q110, the resistor R110, and the MOS transistor M110, these elements constitute a grounded-emitter amplifier in such a configuration that the NPN transistor Q110 and the NPN transistor Q111 share the same base. Therefore, a current i₁₁₁ is equal to a current i₁₁₀ flowing through the NPN transistor Q110. Similarly, currents i₁₁₂, i₁₁₃, and i₁₁₄ flowing through the corresponding NPN transistors Q112, Q113, and Q114 are all equal to the current i₁₁₀ flowing through the NPN transistor Q110. Similarly, currents i₁₂₁, i₁₂₂, i₁₂₃, and i₁₂₄ flowing through the corresponding NPN transistors Q121, Q122, Q123, and Q124 are all equal to the current i₁₂₀ flowing through the NPN transistor Q120.

Further, the currents i₁₁ and i₁₂ flowing through the resistors R31 and R32 in the current-to-voltage converter 14 are given by the following Equations (10) and (11) respectively: i ₁₁ =i ₁₁₁ +i ₁₁₂ +i ₁₁₃ +i ₁₁₄=4×i ₁₁₀=4×(i ₀ −ΔV _(i) /r ₁₀)  (10) i ₁₂ =i ₁₂₁ +i ₁₂₂ +i ₁₂₃ +i ₁₂₄=4×i ₁₂₀=4×(i ₀ +ΔV _(i) /r ₁₀)  (11)

In this case, if resistances of the resistors R31 and R32 are both supposed to be r30, an output signal of the voltage difference ΔV_(o) outputted from the output terminals 19 and 20 of the current-to-voltage converter 14 are given by the following Equation (12): ΔV _(o)=(V _(cc) −r ₃₀ ×i ₁₁)−(V _(cc) −r ₃₀ ×i ₁₂)=r ₃₀×(i ₁₂ −i ₁₁)=4×r ₃₀×{(i ₀ +ΔV _(i)/r₁₀)−(i ₀ −ΔV _(i) /r ₁₀)}=4×2×r ₃₀ ×ΔV _(i) /r ₁₀  (12)

That is, a gain G_(v) of the variable gain amplifier 11 in this condition is given as follows: G_(v)=4×2×r₃₀/r₁₀=4×G₀  (13) Here G₀=2×r₃₀/r₁₀.

Then, if the gain control voltage V_(GC) applied to the input terminal 18 decreases to a level of the voltage V_(b) as shown in FIG. 3, the voltage V_(S1) that appears at the control terminal S1 decreases to 0 V. Accordingly, the N type MOS transistors M111 and M121 are turned off. On the other hand, the N type MOS transistors M112-M114 and M122-M124 remain in the on-state because the voltages at the control terminals S2, S3, and S4 are larger than V_(on). In this case, since the N type MOS transistors M111 and M121 are turned off and a relationship of i₁₁₁=i₁₂₁₌₀ is established, the currents i₁₁ and i₁₂ flowing through the resistors R31 and R32 in the current-to-voltage converter 14 are given by the following Equations (14) and (15), respectively: i ₁₁ =i ₁₁₂ +i ₁₁₃ +i ₁₁₄=3 ×i ₁₁₀=3×(i ₀ −ΔV _(i) /r ₁₀)  (14) i ₁₂ =i ₁₂₂ +i ₁₂₃ +i ₁₂₄=3 ×i ₁₂₀=3×(i ₀ +ΔV _(i) /r ₁₀)  (15)

That is, the gain G_(v) of the variable gain amplifier 11 in this condition is given by: G _(v)=3×2×r ₃₀ /r ₁₀₌₃ ×G ₀  (16)

Similarly, if the gain control voltage V_(GC) applied to the input terminal 18 decreases to a level of the voltage V_(c) as shown in FIG. 3, the N type MOS transistors M112 and M122 are turned off, so that the gain G_(v) of the variable gain amplifier 11 is given by: G _(v)=2×G ₀  (17)

If the gain control voltage V_(GC) applied to the input terminal 18 decreases to a level of the voltage V_(d) as shown in FIG. 3, the N type MOS transistors M113 and M123 are turned off, so that the gain G_(v) of the variable gain amplifier 11 is given by: G _(v)=1×G ₀  (18)

If the gain control voltage V_(GC) applied to the input terminal 18 decreases to a level of the voltage V_(e) as shown in FIG. 3, the N type MOS transistors M114 and M124 are turned off, so that the gain G_(v) of the variable gain amplifier 11 is given by: G_(v)=0×G ₀  (19)

As can be seen from the above, in the variable gain amplifier 11, as the gain control voltage V_(GC) changes, the gain G_(v) changes to 4G₀, 3G₀, 2G₀, G₀, and 0 in this order.

The following describes a case where the gain control voltage V_(GC) is between the voltage V_(a) and the voltage V_(b) shown in FIG. 3, for example. In this case, at the control terminal S1, a voltage between V_(on) and V_(off) appears, so that the N type MOS transistors M111 and M121 whose gates are connected to the control terminal S1 enter an intermediate state between the on-state and the off-state. In this condition, if it is supposed that turn-on resistances of the N type MOS transistors M111 and M121 are R_(ON1), turn-on resistances of the N type MOS transistors M110 and M120 are R_(ON0), base-emitter voltages of the NPN transistors Q110 and Q120 are V_(BE0), and base-emitter voltages of the NPN transistors Q111 and Q121 are V_(BE1), the following Equation (20) is established: V _(BE0)+(R ₁₁₀ +R _(ON0))×i ₁₁₀ =V _(BE1)+(R ₁₁₁ +R _(ON1))×i ₁₁₁  (20)

Equation (20) may be modified into the following Equation (21): V _(BE0) −V _(BE1)+(R ₁₁₀ +R _(ON0))×i ₁₁₀−(R ₁₁₁ +R _(ON1))×i ₁₁₁=0  (21)

Here, given that R₁₁₀>>R_(ON0) and V_(BE0)−V_(BE1)<<(R₁₁₁+R_(ON1))×i₁₁₁−(R₁₁₀+R_(ON0))×i₁₁₀, Equation (21) may be modified into: R ₁₁₀ ×i ₁₁₀−(R ₁₁₁ +R _(ON1))×i ₁₁₁=0  (22)

Since a relationship of R₁₁₀=R₁₁₁ is established, Equation (22) may be modified into: i ₁₁₁ =i ₁₁₀/{1+(R _(ON1) /R ₁₁₁)}  (23)

In a case where the gain control voltage V_(GC) is between the voltages V_(a) and V_(b), the turn-on resistance R_(ON1) of the N type MOS transistors M111 and M121 connected to the control terminal S1 changes in a range of 0 through ∞ in accordance with a gate-source voltage of the N type MOS transistors M111 and M121 that changes between V_(off) and V_(on). Accordingly, the value of the current i₁₁₁ changes in a range of i₁₁₀ to 0. Further, the following equations (24) and (25) are established: i ₁₁ =i ₁₁₁ +i ₁₁₂ +i ₁₁₃ +i ₁₁₄₌₃ ×i ₁₁₀ +i ₁₁₀/{1+R _(ON1) /R ₁₁₁}  (24) i ₁₂ =i ₁₂₁ +i ₁₂₂ +i ₁₂₃ +i ₁₂₄₌₃ ×i ₁₂₀ +i ₁₂₀/{1+R _(ON1) /R ₁₂₁}  (25)

Thus, the gain G_(v) is given by the following Equation (26): G _(v)=3×G ₀+{1+(R _(ON1) /R ₁₂₁)}×G ₀  (26)

As seen from the above, in the variable gain amplifier 11 according to the second embodiment, the gain G_(v) changes from 4G₀ to 3G₀ continuously. Similarly, in cases where the gain control voltage V_(GC) is between the voltage V_(b) and the voltage V_(c), between the voltage V_(c) and the voltage V_(d), and between the voltage V_(d) and V_(e), the gain G_(v) continuously changes from 3G₀ to 2G₀, from 2G₀ to G₀, and from G₀ to 0, respectively, so that the gain G_(v) eventually changes from 0 to 4G₀ continuously.

FIG. 7 is a graph showing a relationship between the gain control voltage V_(GC) and the gain G_(v) for the variable gain amplifier 11 according to the second embodiment. As shown in FIG. 7, the gain G_(v), continuously changes from 0 to 4G₀ as the gain control voltage V_(GC) changes.

The graph shown in FIG. 7 indicates a poor linearity, that is, a poor linearity of a change in gain G_(v) with respect to the gain control voltage V_(GC), which poor linearity can be solved by increasing the number of the current amplification circuits of the current amplifier 13. Although the variable gain amplifier 11 shown in FIG. 2 has the four current amplification circuits 31, 32, 33, and 34 in the current amplifier 13, this number can be increased to, for example, 10. Accordingly, also the number of the corresponding control terminals of the control voltage generator 15 can be increased to 10, to improve the gain linearity. FIG. 8 is a graph showing a relationship between the gain control voltage V_(GC) and the gain G_(v) in a case where the number of the current amplification circuits is increased to 10. As shown in FIG. 8, by increasing the number of the current amplifiers, the linearity of the gain G_(v) can be improved.

Furthermore, by changing the gain control voltage VGC continuously, the current i₁₁ flowing through the resistor R31 changes continuously. FIG. 9 is a graph showing a relationship between the gain control voltage V_(GC) and the current i₁₁. Comparison between FIGS. 8 and 9 shows that if the gain G_(v) is high, the current i₁₁ is large, and if the gain G_(v) is low, the current is small. This relationship is similarly established between the gain control voltage V_(GC) and the current i₁₂ flowing through the resistor R32. That is, the variable gain amplifier 11 according to the second embodiment can satisfy a feature that dissipation power is low if the gain is low and an output level is low.

Further, the saturation input level, which is r₁₀×i₀, of the variable gain amplifier 11 according to the second embodiment does not change even if the gain changes. That is, the variable gain amplifier 11 according to the second embodiment can satisfy a feature that the saturation input level does not change even if the gain changes.

In the variable gain amplifier 11 according to the present invention, although the current-to-voltage converter 14 is constituted of the resistors R31 and R32, it may be constituted of an inductor and a capacitor. The following describes a difference between the case where the current-to-voltage converter 14 is constituted of a resistor and the case where it is constituted of an inductor. In the case where the current-to-voltage converter 14 is constituted of the resistors R31 and R32, if a resistance of the resistor R31 is assumed to be r₃₀, the voltage amplitude V_(o1) at the output terminal 19 is given by the following Equation (27): V _(o1) =V _(cc) −r ₃₀ ×i ₁₁  (27)

If it is assumed here that, when ΔV_(i)=0 is established, i_(11a) is i_(11a) and that V_(o1a)=V_(cc)−r₃₀×i_(11a) and i₁₁=i_(11a)+Δi₁₁, the following Equation (28) is established: $\begin{matrix} \begin{matrix} {V_{o\quad 1} = {V_{cc} - {r_{30} \times \left( {i_{11a} + {\Delta\quad i_{11}}} \right)}}} \\ {= {V_{cc} - {r_{30} \times i_{11a}} - {r_{30} \times \Delta\quad i_{11}}}} \\ {= {V_{o\quad 1a} - {r_{30} \times \Delta\quad i_{11}}}} \end{matrix} & (28) \end{matrix}$

Since Δi₁₁ fluctuates with ΔV_(i), it can be known from Equation (28) that an amplitude of the voltage V_(o1) fluctuates around the voltage V_(o1a).

It is to be noted that a lower limit of the voltage V_(o1) is a voltage Vsat around at which none of the NPN transistors Q110-Q114 is saturated, and its upper limit is the power source voltage V_(cc). That is, if the current-to-voltage converter 14 is constituted of a resistor, a relationship of V_(sat)<V_(o1)<V_(cc) is established. Similarly, a relationship Of V_(sat)<V_(o2)<V_(cc) is established also with the amplitude of the voltage V_(o2) at the output terminal 20.

Next, the case is discussed where this current-to-voltage converter 14 is constituted of an inductor. FIG. 10 is a circuit diagram showing a configuration example of the case where the current-to-voltage converter 14 is constituted of an inductor. In the current-to-voltage converter 14 shown in FIG. 10, the resistors R31 and R32 in the current-to-voltage converter 14 shown in FIG. 2 are replaced by inductors L31 and L32. If an impedance of the inductor L31 is assumed to be Z_(L)=j×ω×L, an amplitude of the voltage V_(o1) at the output terminal 19 is given by the following Equation (29): $\begin{matrix} \begin{matrix} {V_{o\quad 1} = {V_{cc} - {Z_{L} \times i_{11}}}} \\ {= {V_{cc} - {Z_{L} \times i_{11a}} - {Z_{L} \times \Delta\quad i_{11}}}} \end{matrix} & (29) \end{matrix}$

Since i_(11a) in this equation is a current component having no frequency fluctuations, Z_(L)×i_(11a)=0 is given, and thus the following Equation (30) is established: V _(o1) =V _(cc) −Z _(L) ×Δi ₁₁  (30)

That is, it can be known that the amplitude of the voltage V_(o1) fluctuates around V_(cc). Furthermore, since Z_(L)×Δi₁₁ may take on a negative value in some cases, V_(o1)>V_(cc) is established. Further, similarly, V_(o2)>V_(cc) is established also with the amplitude of the voltage V_(o2) at the output terminal 20.

FIGS. 11 and 12 are graphs describing the above-described contents. Specifically, FIG. 11 is a graph showing a relationship between a voltage V_(o1) at the output terminal 19 and the current i₁₁ flowing through the resistor R31 in the case where the current-to-voltage converter 14 is constituted of the resistors R31 and R32. FIG. 12 is a graph showing a relationship between the voltage V_(o1) at the output terminal 19 and the current i₁₁ flowing through the inductor L31 in the case where the current-to-voltage converter 14 is constituted of the inductors L31 and L32. In these graphs, a vertical axis represents the voltage V_(o1) at the output terminal 19 and a horizontal axis represents the current i₁₁. As can be seen from comparison between FIGS. 11 and 12, in the case where the current-to-voltage converter 14 is constituted of an inductance, the voltage V_(o1) at the output terminal 19 has a larger amplitude than that in the case where it is constituted of a resistor, thus resulting in Z_(L)>r₃₀. That is, the gain G_(v) of the variable gain amplifier 11 can be increased.

Further, the current-to-voltage converter 14 may be constituted of an inductor and a capacitor connected in parallel with each other. FIG. 13 is a circuit diagram showing a configuration example of the current-to-voltage converter 14. In the current-to-voltage converter 14 shown in FIG. 13, the resistor R31 and the resistor R32 in the current-to-voltage converter 14 shown in FIG. 2 are replaced by the inductor L31 and a capacitor C31 connected in parallel with each other and the inductor 132 and a capacitor C32 connected in parallel with each other, respectively. In the case where the current-to-voltage converter 14 is configured as shown in FIG. 13, an impedance ZLC of the parallel connection of the inductor and the capacitor is Z_(LC)=j×ω×L/(1−ω²×L×C). Therefore, this impedance Z_(LC) takes on a very large value if ω approaches a value of a square root of L×C, that is, it is in the vicinity of ω=√{square root over ((L×C))}. Accordingly, if the current-to-voltage converter 14 is constituted of a combination of an inductor and a capacitor, the gain G_(v) of the variable gain amplifier 11 can be increased more than that in a case where it is constituted of a standalone resistor or a standalone inductor.

It is to be noted that the voltage-to-current converter 12 can be modified to another configuration. FIG. 14 is a circuit diagram showing another configuration example of the voltage-to-current converter 12. As shown in FIG. 14, the voltage V_(i1) of an input signal to the input terminal 16 is supplied to a non-inverting input terminal (+) of the operational amplifier 21 whose output voltage is in turn inputted to a base of a PNP transistor Q151. A collector of the PNP transistor Q151 is connected to the resistor R10 and the constant current source 23, while a collector voltage of the PNP transistor Q151 is fed back to an inverting input terminal (−) of the operational amplifier 21. An emitter of the PNP transistor Q151 is connected to the power source voltage V_(cc). Thus, a voltage of the collector of the PNP transistor Q151 is fed back to the operational amplifier 21 and controlled to be equal to the voltage V_(i1) of the input terminal 16. Similarly, a voltage of a collector of a PNP transistor Q161 is fed back to the operational amplifier 22 and controlled to be equal to the voltage V_(i2) of the input terminal 17.

In this case, Δi flowing through the resistor R10 is expressed in the following Equation (31), provided that ΔV_(i)=V_(i1)−V_(i2): Δi=(V _(i1) −V _(i2))/r ₁₀ =ΔV _(i) /r ₁₀  (31)

In this configuration, if output currents from the constant current sources 23 and 24 are both supposed to be i₀, a current i₁₅₀ that flows through the PNP transistor Q151 and a current i₁₆₀ that flows through the PNP transistor Q161 are given by the following Equations (32) and (33), respectively: i ₁₅₀ =i ₀ −Δi=i ₀ −ΔV _(i) /r ₁₀  (32) i ₁₆₀ =i ₀ −Δi=i ₀ +ΔV _(i) /r ₁₀  (33)

In this case, i₁₅₀>0 and i₁₆₀>0 are established, so that Δi changes in a range of −i₀ through +i₀ in accordance with ΔV_(i). A voltage of ΔV_(i) required for causing Δi to change in the range of −i₀ through +i₀ is r₁₀×i₀. Conversely, even if ΔV_(i) in excess of this voltage is input, Δi cannot follow a change in value of ΔV_(i). Therefore, this voltage provides a saturation input level of this variable gain amplifier. Further, the bases and the emitters of the PNP transistors Q151 and Q152 are commonly connected respectively, and the bases and the emitters of the PNP transistors Q161 and Q162 are commonly connected respectively. Therefore, a current flowing through the PNP transistor Q152 is equal to the current i₁₅₀ flowing through the PNP transistor Q151, and a current flowing through the PNP transistor Q162 is equal to the current i₁₆₀ flowing through the PNP transistor Q161. Furthermore, the collectors of the PNP transistors Q152 and the NPN transistor Q110 are commonly connected, so that the same current flows through the PNP transistor Q152 and the PNP transistor Q110. Further, the collectors of the PNP transistors Q162 and the NPN transistor Q120 are commonly connected, so that the same current flows through the PNP transistor Q162 and the PNP transistor Q120. Thus, it can be known that the voltage-to-current converter 12 shown in FIG. 14 and the voltage-to-current converter 12 shown in FIG. 2 work in the same way. It is to be noted that each of the two configurations shown in FIGS. 14 and 2 respectively can be used depending on a use properly.

Although in the above-described variable gain amplifier 11, to simplify description, it is supposed that the NPN transistors Q110-Q114 and Q120-Q124 have the same size and properties, the resistors R110-R114 and R120-124 have the same resistance, the N type MOS transistors M110-M114 and M120-M124 have the same size and properties, and the resistors R31 and R32 have the same resistance, they may be set as follows.

For example, it is supposed that each of the NPN transistors Q110 and Q120 has a size of 1, each of the NPN transistors Q111 and Q121 has a size of 4, each of the NPN transistors Q112 and Q122 has a size of 2, each of the NPN transistors Q113 and Q123 has a size of 1, and each of the NPN transistors Q114 and Q124 has a size of 1. Further, it is supposed that each of the N type MOS transistors M110 and M120 has a size of 1, each of the N type MOS transistors M111 and M121 has a size of 4, each of the N type MOS transistors M112 and M122 has a size of 2, each of the N type MOS transistors M113 and M123 has a size of 1, and each of the N type MOS transistors M114 and M124 has a size of 1. Further, it is supposed that each of the resistors R110 and R120 has a resistance of r₀, each of the resistors R111 and R121 has a resistance of r₀/4, each of the resistors R112 and R122 has a resistance of r₀/2, each of the resistors R113 and R123 has a resistance of r₀, and each of the resistors R114 and R124 has a resistance of r₀. In this configuration, the current amplification circuits 33 and 34 are in the same conditions as those of the variable gain amplifier 11 shown in FIG. 2, but, in the current amplification circuit 31, the NPN transistors and the N type MOS transistors respectively have four times the size and one-fourth the resistance value of the resistors. Therefore, the configuration is equivalent to the four current amplification circuits 31 integrated in parallel, thereby resulting in four times the current amplification factor of the current amplification circuit 31 shown in FIG. 2. Similarly, the current amplification circuit 32 has twice the current amplification factor of the current amplification circuit 32 shown in FIG. 2.

The following describes an operation of a modification of the variable gain amplifier 11. If the gain control voltage V_(GC) inputted to the input terminal 18 is equal to the voltage V_(a) shown in FIG. 3, the voltages that appear at the output terminals S1-S4 are each larger than the V_(on) sufficiently. In this case, the following Equations (34) and (35) are established: i ₁₁ =i ₁₁₁ +i ₁₁₂ +i ₁₁₃ +i ₁₁₄=4×i ₁₁₀+2×i ₁₁₀ +i ₁₁₀ +i ₁₁₀=8×i ₁₁₀  (34) i ₁₂ =i ₁₂₁ +i ₁₂₂ +i ₁₂₃ +i ₁₂₄=4×i ₁₂₀+2×i ₁₂₀ +i ₁₂₀ +i ₁₂₀=8×i ₁₂₀  (35)

From these Equations (34) and (35), the following Equation (36) is given: G _(v)=8×G ₀  (36)

Next, if the gain control voltage V_(GC) inputted to the input terminal 18 is equal to the voltage V_(b) shown in FIG. 3, the N type MOS transistors M111 and M121 are turned off, so that the currents i₁₁₁ and i₁₂₁ are 0. Therefore, the following Equations (37) and (38) are established: i ₁₁ =i ₁₁₁ +i ₁₁₂ +i ₁₁₃ +i ₁₁₄₌₀ ×i ₁₁₀+2×i ₁₁₀ +i ₁₁₀ +i ₁₁₀=4×i ₁₁₀  (37) i ₁₂ =i ₁₂₁ +i ₁₂₂ +i ₁₂₃ +i ₁₂₄₌₀ ×i ₁₂₀+2×i ₁₂₀ +i ₁₂₀ +i ₁₂₀=4×i ₁₂₀  (38)

From these Equations (37) and (38), the following Equation (39) is given: G _(v)=4×G ₀  (39)

Similarly, if the gain control voltage V_(GC) input to the input terminal 18 is equal to the voltage V_(c) shown in FIG. 3, the gain G_(v) is given by: Equation (40) G _(v)=2×G ₀  (40) and if the gain control voltage V_(GC) input to the input terminal 18 is equal to the voltage V_(d) shown in FIG. 3, the gain G_(v) is given by: Equation (41) G _(v)=1×G ₀  (41)

In such a manner, as the gain control voltage V_(GC) decreases, the gain G_(v) changes to 8G₀, 4G₀, 2G₀, and G₀ in this order. A decibel gain calculated on the assumption that 20LOG(G₀)=GdB₀ changes to 18+ GdB₀, 12+ GdB₀, 6+ GdB₀, and GdB₀ in this order as the gain control voltage V_(GC) decreases. From this, it can be seen that, as the gain control voltage V_(GC) changes, the decibel gain changes linearly in a step of 6 dB, that is, LOG(2)dB. By thus setting the current amplification factors of the current amplification circuits 31, 32, 33, and 34 to be 4, 2, 1, and 1 respectively, it is possible to change the decibel gain linearly in a step of 6 dB.

Similarly, if the current amplification factors of the current amplification circuits 31, 32, 33, and 34 are set to 18, 6, 2, and 1 respectively, the gain G_(v) changes to 27G₀, 9G₀, 3G₀, and G₀ in this order and the decibel gain changes to 28.6+ GdB₀, 19.1+ GdB₀, 9.5+ GdB₀, and GdB₀ in this order. From this, it can be seen that the decibel gain changes linearly in a step of 9.5 dB, that is, 20LOG(3)dB. Although the current amplification factors of the current amplification circuits 31, 32, 33, and 34 have been respectively set to 18, 6, 2, and 1 as an example, they may be set to 9, 3, 1, and 0.5 or to 36, 12, 4, and 2, respectively.

It is to be noted that if N and n are supposed to be an integer of 2 or larger, N number of the current amplification circuits are provided, so that by setting a current amplification factor N₁ of the first one of these current amplification circuits to α₀ and a current amplification factor Nn of the n-th one of these current amplification circuits to N_(n)=α₀×(A−1)×A^((n−2)), it is possible to linearly change the decibel gain of the variable gain amplifier 11 in a step of 20LOG(A).

Further, although, as for voltage values V_(b), V_(c), V_(d), V_(e) of the gain control voltage V_(GC) shown in FIG. 3, the voltage differences V_(b)−V_(c), V_(c)−V_(d), and V_(d)−V_(e) are supposed to be equal to each other, they need not necessarily be equal to each other; by adjusting these voltage differences, and it is possible to improve the linearity of a change in gain with respect to the gain control voltage.

Further, of course, the variable gain amplifier 11 according to the second embodiment can also satisfy the three features of low dissipation power, low distortion, and gain continuity similarly to the variable gain amplifier 1 according to the first embodiment.

A variable gain amplifier according to the present invention can be used in a mixer or a quadrature modulator. The following describes a mixer and a quadrature modulator using a variable gain amplifier according to the present invention. FIG. 15 is a circuit diagram showing a mixer that uses the variable gain amplifier 11 according to the present invention. As shown in FIG. 15, a mixer 61 includes a variable gain amplifier that includes the voltage-to-current converter 12, current amplifier 13, current-to-voltage converter 14, and control voltage generator 15, and further includes a Gilbert cell section 62 which constitutes a mixing unit. The voltage-to-current converter 12, the current amplifier 13, the current-to-voltage converter 14, and the control voltage generator 15 have the same configurations as those shown in FIG. 2, and the description thereof is omitted. The external voltages V_(i1) and V_(i2) having a voltage difference ΔV_(i) are input to the input terminals 16 and 17 of the voltage-to-current converter 12. Meanwhile, conversion gain control signals Lo1 and Lo2 having a voltage difference ΔV_(Lo) are externally applied to input terminals 63 and 64 of the Gilbert cell section 62, and the output signals having a voltage difference ΔV_(o) are obtained from the output terminals 19 and 20 of the current-to-voltage converter 14 as a gain-controlled signal by multiplying the voltage differences ΔV_(i) and ΔV_(Lo).

The Gilbert section 62 includes four NPN transistors Q171-Q174. Bases of the NPN transistors Q171 and Q174 are connected commonly and bases of the NPN transistors Q172 and Q173 are connected commonly. A voltage inputted to the input terminal 63 is applied to the bases of the NPN transistors Q171 and Q174, and a voltage inputted to the input terminal 64 is applied to the bases of the NPN transistors Q172 and Q173. Further, collectors of the NPN transistors Q171 and Q173 are connected commonly and collectors of the NPN transistors Q172 and Q174 are connected commonly. The resistor R31 of the current-to-voltage converter 14 is connected to the collectors of the NPN transistors Q171 and Q173, and the resistor R32 of the current-to-voltage converter 14 is connected to the collectors of the NPN transistors Q172 and Q174.

The following describes an operation of the mixer 61. If the voltage difference ΔV_(Lo) between the input voltages Lo1 and Lo2 at the input terminals 63 and 64 of the Gilbert cell section 62 is properly large, the NPN transistors Q171-Q174 in the Gilbert cell section 62 operate as simple current-transfer switches controlled by the voltage difference ΔV_(Lo). That is, if the input terminals 63 and 64 of the Gilbert cell section 62 are supplied with a properly large voltage difference ΔV_(Lo) and a voltage applied to the input terminal 63 is larger than a voltage applied to the input terminal 64 (ΔV_(Lo)>0), the NPN transistors Q171 and Q174 are turned on and the NPN transistors 172 and 173 are turned off. In this case, currents i₃₁ and i₃₂ flowing through the resistors R31 and R32 in the current-to-voltage converter 14 respectively are i₃₁=i₁₁ and i₃₂=i₁₂, so that if R31=R32=r₃₀ is assumed, the output voltage difference ΔV_(o) obtained from the output terminals 19 and 20 of the current-to-voltage converter 14 is given by the following Equation (42): ΔV _(o) =r ₃₀×(i ₃₂ −i ₃₁)=r ₃₀×(i ₁₂ −i ₁₁)  (42)

On the other hand, if a voltage applied to the input terminal 63 of the Gilbert cell section 62 is smaller than a voltage applied to the input terminal 64 (i.e., ΔV_(Lo)<0), the NPN transistors Q171 and Q174 are turned off and the NPN transistors 172 and 173 are turned on. In this case, the currents i₃₁ and i₃₂ flowing through the resistors R31 and R32 in the current-to-voltage converter 14 respectively are i₃₁=i₁₂ and i₃₂=i₁₁, so that the output voltage difference ΔV_(o) obtained from the output terminals 19 and 20 of the current-to-voltage converter 14 is given by the following Equation (43): ΔV _(o) =r ₃₀×(i ₃₂ −i ₃₁)=r ₃₀×(i ₁₁ −i ₁₂)  (43)

That is, the output voltage difference ΔV_(o) has the same frequency as the input voltage difference ΔV_(Lo) and is given by multiplying a square wave having an amplitude of ±1 and r₃₀×(i₁₂−i₁₁). If the gain G_(v) of the variable gain amplifier 11 according to the second embodiment is applied in this case, the voltage difference can be given by Equations (44) and (45) as follows: ΔV _(o) =G _(v) ×ΔV _(i) (ΔV _(Lo)>0)  (44) ΔV _(o) =−G _(v) ×ΔV _(i) (ΔV _(Lo)>0)  (45)

Accordingly, the gain G_(v) can be controlled by using the gain control voltage V_(GC) inputted to the input terminal 18.

Similarly, in the case of this mixer 61 also, although a current is large if the gain G_(v) is high and the output voltage difference ΔV_(o) is large, the current can be decreased if the gain is low and the output voltage difference ΔV_(o) is small. Therefore, a mixer that uses a variable gain amplifier according to the present invention can satisfy a feature of dissipation power being low if the gain is small and the output level is low. Further, a mixer that uses a variable gain amplifier according to the present invention can satisfy a feature that even if a gain is changed, a saturation input level remains unchanged with distortion being low. Further, a mixer that uses a variable gain amplifier according to the present invention can easily realize linearity of a continuous gain irrespective of a value of the gain control voltage.

Further, in the mixer that uses a variable gain amplifier according to the present invention, it is possible to always switch the NPN transistors Q171-Q174 in the Gilbert cell section 62 while a level of ΔV_(Lo) is unchanged during the gain control. Generally, better distortion and noise characteristics can be obtained by always switching the NPN transistors in the Gilbert section 62 in use. Therefore, a mixer that uses the variable gain amplifier 11 according to the present invention can realize gain control while keeping up good distortion and noise characteristics of the Gilbert cell section 62.

Further, similarly to the variable gain amplifier 11 according to the second embodiment, by constituting the current-to-voltage converter 14 of an inductor rather than a resistor, an amplitude of a voltage at the output terminals 19 and 20 can be increased, thereby increasing the gain G_(v) of the mixer 61. Further, by constituting the current-to-voltage converter 14 of an inductor and a capacitor connected in parallel to each other, the gain G_(v) of the mixer 61 can be increased.

Further, the above-described mixers 61 can be combined to realize a quadrature modulator. FIG. 16 shows a block diagram showing a configuration example of a quadrature modulator 71 that uses a variable gain amplifier according to the present invention. As shown in FIG. 16, the quadrature modulator 71 includes voltage-to-current converters 12 a and 12 b, current amplifiers 13 a and 13 b, a control voltage generator 15, mixing units 62 a and 62 b, a current-to-voltage converter 14, and an adder 72. Each of the mixing units 62 a and 62 b corresponds to the Gilbert cell section 62 in the mixer 61. The configurations of the voltage-to-current converters 12 a and 12 b, current amplifiers 13 a and 13 b, control voltage generator 15, mixing units 62 a and 62 b, current-to-voltage converter 14 have already been described. A voltage V_(in) inputted to input terminals 16 a and 17 a of the voltage-to-current converter 12 a is mixed in accordance with a conversion gain control signal Lo1 inputted to input terminals 63 a and 64 a of the mixing unit 62 a. Meanwhile, a voltage V_(inb) inputted to input terminals 16 b and 17 b of the voltage-to-current converter 12 b is mixed in accordance with a conversion gain control signal Lo2 inputted to input terminals 63 b and 64 b of the mixing unit 62 b. Output signals of the mixing units 62 a and 62 b are added up by the adder 72 and the added signal is converted by the current-to-voltage converter 14 to provide output voltage signals. It is to be noted here that the output signals Lo2 and Lo1 have the same frequency and amplitude and different phases shifted by 90 degrees from each other.

In this quadrature modulator 71, although a current is large if the gain G_(v) is high and the output voltage difference ΔV_(o) is large, the current can be decreased if the gain is low and the output voltage difference ΔV_(o) is small. Therefore, the quadrature modulator that uses a variable gain amplifier according to the present invention can satisfy a feature of dissipation power being low if the gain is small and the output level is low. Further, the quadrature modulator that uses a variable gain amplifier according to the present invention can satisfy a feature that even if a gain is changed, a saturation input level remains unchanged with distortion being low. Further, the quadrature modulator that uses a variable gain amplifier according to the present invention can easily realize linearity of a continuous gain irrespective of a value of the gain control voltage.

Further, when using a quadrature modulator that uses a variable gain amplifier according to the present invention, it is possible to always switch the NPN transistors in the mixing units 62 a and 62 b while a level of ΔV_(Lo) remains unchanged during the gain control operation. Generally, better distortion and noise characteristics can be obtained by always switching the NPN transistors in these mixing units 62 a and 62 b in use. Therefore, a quadrature modulator that uses the variable gain amplifier 11 according to the present invention can realize the gain control while keeping up good distortion and noise characteristics of the mixing units 62 a and 62 b.

A variable gain amplifier according to the present invention can satisfy the three features of lower power dissipation, low distortion, and gain continuity simultaneously and it is useful as an amplifier to be used in a transmission section of, for example, a CDMA wireless terminal. 

1. A variable gain amplifier comprising: a voltage-to-current converter for converting an input voltage into a current; a current amplifier for amplifying the current converted by the voltage-to-current converter; a current-to-voltage converter for converting the current amplified by the current amplifier into a voltage and outputting the resulting voltage; and a controller for controlling a gain of the variable gain amplifier by varying an amplification factor of the current amplifier in accordance with a gain control signal.
 2. The variable gain amplifier according to claim 1, wherein the current amplifier has a current mirror circuit which inputs the current converted by the voltage-to-current converter, and the controller varies a ratio of an output current to an input current of the current mirror circuit in accordance with the gain control signal.
 3. The variable gain amplifier according to claim 2, wherein the current mirror circuit includes an input-side transistor through which the current converted by the voltage-to-current converter flows; an output-side transistor through which a current proportional to the current flowing through the input-side transistor flows; and a variable resistor connected in series to the output-side transistor, and wherein the current-to-voltage converter converts the current flowing through the output-side transistor into a voltage and outputs the resulting voltage, and the controller varies a resistance of the variable resistor in accordance with the gain control signal.
 4. The variable gain amplifier according to claim 3, wherein the variable resistor is constituted of an MOS transistor, and the controller varies a control voltage of the MOS transistor in accordance with the gain control signal.
 5. The variable gain amplifier according to claim 1, wherein the current amplifier has a plurality of current amplification circuits each amplifying the current converted by the voltage-to-current converter and outputting the amplified current, and the controller varies an amplification factor of each of the current amplification circuits in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents outputted from the current amplification circuits into a voltage.
 6. The variable gain amplifier according to claim 5, wherein the voltage-to-current converter has a pair of first and second transistors through each of which a current converted from the input voltage flows, and each of the current amplification circuits has a pair of third and fourth transistors to constitute a group of third transistors and a group of fourth transistors in the current amplifier, wherein base electrodes of the group of the third transistors are commonly connected to the base of the first transistor and collector electrodes of the group of the third transistors are commonly connected so that a current proportional to a current flowing through the first transistor flows through each of the group of the third transistors, and base electrodes of the group of the fourth transistors are commonly connected to the base of the second transistor and collector electrodes of the group of the fourth transistors are commonly connected so that a current proportional to a current flowing through the second transistor flows through each of the group of the fourth transistors, and wherein the controller varies a quantity of a current flowing through each of the third and fourth transistors in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents flowing through the group of the third transistors and the group of the fourth transistors into a voltage.
 7. The variable gain amplifier according to claim 6, wherein each of the current amplification circuits has a variable resistor connected in series to each of the third transistors and a variable resistor connected in series to each of the fourth transistors, and the controller varies a resistance of each of the variable resistors in accordance with the gain control signal.
 8. The variable gain amplifier according to claim 6, wherein each of the current amplification circuits has a MOS transistor connected in series to each of the third transistors and a MOS transistor connected in series to each of the fourth transistors, and the controller varies a control voltage of each of the MOS transistors in accordance with the gain control signal, thereby varying a quantity of a current flowing through each of the third and fourth transistors.
 9. The variable gain amplifier according to claim 8, wherein the controller has a constant voltage generator that generates a constant voltage to be applied to each of the current amplification circuits and a plurality of control voltage generators each of which generates and outputs a control voltage to be applied to each pair of the MOS transistors in each of the current amplification circuits, and wherein each of the control voltage generators has a differential amplification circuit to which a constant voltage generated by the constant voltage generator and a voltage of the gain control signal are inputted to the input terminals thereof, respectively, a current mirror circuit whose input-side transistor constitutes a load of one of a couple of differential transistors of the differential amplification circuit, and a voltage output section for converting an output current of the current mirror circuit into a voltage and outputting the resultant converted voltage.
 10. The variable gain amplifier according to claim 1, wherein the current-to-voltage converter is comprised of a resistor or an inductor.
 11. The variable gain amplifier according to claim 1, wherein the current-to-voltage converter has an inductor and a capacitor connected in parallel to each other.
 12. A mixer which converts a frequency of an input voltage in accordance with a conversion gain control signal supplied from an outside and outputs the resulting voltage having a frequency converted, said mixer comprising: a voltage-to-current converter for converting the input voltage into a current; a current amplifier for amplifying the current converted by the voltage-to-current converter; a controller for controlling an amplification factor of the current amplifier in accordance with a gain control signal; a mixing unit for converting the frequency of the current amplified by the current amplifier, in accordance with the conversion gain control signal; and a current-to-voltage converter for converting the resulting frequency-converted current into a voltage.
 13. The mixer according to claim 12, wherein the current amplifier has a current mirror circuit which inputs the current converted by the voltage-to-current converter, and the controller varies a ratio of an output current to an input current of the current mirror circuit in accordance with the gain control signal.
 14. The mixer according to claim 13, wherein the current mirror circuit includes an input-side transistor through which the current converted by the voltage-to-current converter flows; an output-side transistor through which a current proportional to the current flowing through the input-side transistor flows; and a variable resistor connected in series to the output-side transistor, and wherein the current-to-voltage converter converts the current flowing through the output-side transistor into a voltage and outputs the resulting voltage, and the controller varies a resistance of the variable resistor in accordance with the gain control signal.
 15. The mixer according to claim 14, wherein the variable resistor is constituted of a MOS transistor and the controller varies a control voltage of the MOS transistor in accordance with the gain control signal.
 16. The mixer according to claim 12, wherein the current amplifier has a plurality of current amplification circuits each amplifying the current converted by the voltage-to-current converter and outputting the amplified current, and wherein the controller varies an amplification factor of each of the current amplification circuits in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents outputted from the current amplification circuits into a voltage.
 17. The mixer according to claim 16, wherein the voltage-to-current converter has a pair of first and second transistors through each of which a current converted from the input voltage flows, and each of the current amplification circuits has a pair of third and fourth transistors to constitute a group of third transistors and a group of fourth transistors in the current amplifier, wherein base electrodes of the group of the third transistors are commonly connected to the base of the first transistor and collector electrodes of the group of the third transistors are commonly connected so that a current proportional to a current flowing through the first transistor flows through each of the group of the third transistors, and base electrodes of the group of the fourth transistors are commonly connected to the base of the second transistor and collector electrodes of the group of the fourth transistors are commonly connected so that a current proportional to a current flowing through the second transistor flows through each of the group of the fourth transistors, and wherein the controller varies a quantity of a current flowing through each of the third and fourth transistors in accordance with the gain control signal, and the current-to-voltage converter converts a total sum of currents flowing through the group of the third transistors and the group of the fourth transistors into a voltage.
 18. The mixer according to claim 17, wherein each of the current amplification circuits has a variable resistor connected in series to each of the third transistors and a variable resistor connected in series to each of the fourth transistors, and the controller varies a resistance of each of the variable resistors in accordance with the gain control signal.
 19. The mixer according to claim 17, wherein each of the current amplification circuits has a MOS transistor connected in series to each of the third transistors and a MOS transistor connected in series to each of the fourth transistors, and the controller varies a control voltage of each of the MOS transistors in accordance with the gain control signal, thereby varying a quantity of a current flowing through each of the third and fourth transistors.
 20. The mixer according to claim 19, wherein the controller has a constant voltage generator that generates a constant voltage to be applied to each of the current amplification circuits and a plurality of control voltage generators each of which generates and outputs a control voltage to be applied to each pair of the MOS transistors in each of the current amplification circuits, and wherein each of the control voltage generators has a differential amplification circuit to which a constant voltage generated by the constant voltage generator and a voltage of the gain control signal are inputted to the input terminals thereof, respectively, a current mirror circuit whose input-side transistor constitutes a load of one of a couple of differential transistors of the differential amplification circuit, and a voltage output section for converting an output current of the current mirror circuit into a voltage and outputting the resultant converted voltage.
 21. The mixer according to claim 12, wherein the current-to-voltage converter is comprised of a resistor or an inductor.
 22. The mixer according to claim 12, wherein the current-to-voltage converter has an inductor and a capacitor connected in parallel to each other.
 23. A quadrature modulator which includes a pair of mixers, an adder for adding up output currents of the mixers, and a current-to-voltage converter for converting an output current of the adder into a voltage, wherein each of the mixers comprises: a voltage-to-current converter for converting an input voltage into a current; a current amplifier for amplifying the current converted by the voltage-to-current converter; a controller for controlling an amplification factor of the current amplifier in accordance with a gain control signal; and a mixing unit for converting a frequency of the current amplified by the current amplifier, in accordance with a conversion gain control signal. 